The present invention relates in general to communication systems and components, and is particularly directed to a circuit arrangement for controllably adjusting DC biasing and overhead voltage characteristics for a subscriber line interface circuit (SLIC), in a manner that is optimized for the mode of operation of the SLIC.
In order to facilitate interfacing with a variety of telecommunication circuits, including those providing digital codec functionality, the subscriber line interface circuits, or SLICs, employed by telecommunication service providers must comply with a demanding set of performance requirements including accuracy, linearity, insensitivity to common mode signals, filtering, low power dissipation, low noise, and ease of impedance matching programmability. In addition, for different installations, the length of the wireline pair to which the SLIC is connected is not only expected to vary, but may be very significant (e.g., on the order of multiple miles); this wireline pair transports both substantial DC voltages, as well as AC signals (e.g., voice and/or ringing). As a consequence, it has been difficult to realize a SLIC implementation that has xe2x80x98universalxe2x80x99 use in both legacy and state of the art applications.
Advantageously, the SLIC transmission channel described in the above-referenced ""505 application effectively realizes these objectives by the combination of a front end, current-sensing transimpedance stage coupled in cascade with a transconductance amplifier-configured filter/gain output stage. The front end transimpedance stage is coupled to respective tip and ring portions of a telecommunication wireline pair, and is operative to transform differentially sensed tip and ring input currents into a precise, single ended voltage. This voltage is converted by the transconductance amplifier-based filter/gain output stage into a very precise, single ended output current, which is then transformed into as a single ended output voltage for application to a current-sense, voltage feed-feed telecommunication circuit. In addition, the transmission channel of the ""505 application is configured to have its passband for AC signals programmable by means of a single external reactance component (capacitor); also, the output impedance it presents to the line is programmable by means of only one programming pin.
Because a SLIC is required to perform a variety of signal-coupling and conditioning tasks, including DC biasing of an associated telephone circuit, as well as providing the appropriate overhead voltage on the wireline pair, as described above, design of the SLIC is critical to power management and signal transport fidelity. For example, too low an overhead voltage across the tip and ring pair can result in insufficient signal amplitude headroom, which may lead to unwanted clipping of the AC (voice) signals being transmitted through the subscriber loop. On the other hand, an excessive overhead voltage could result in not enough current to bias a telephone connected to a long loop.
In accordance with the present invention, these concerns are successfully addressed by means of a new and improved DC biasing circuit architecture that is configured to controllably set the differential DC voltage characteristic of a tip-ring loop driven by a subscriber line interface circuit (SLIC), in a manner that is optimized for the mode of operation of the telephone circuit. Such modes of operation include on-hook, open circuit mode, in which the subscriber""s phone is disconnected; on-hook, quiescent mode, in which the SLIC is minimally active, such as for the purpose of monitoring the line for a data transmission (such as caller ID); transition mode, where the subscriber is in the process of going off-hook and placing a call; and off-hook mode, where the SLIC is in its active call (voice transmission) mode. These respective modes of operation require different dynamic ranges of overhead voltage, while supplying the current required for proper biasing of the wireline pair.
Pursuant to the invention, the DC feed characteristics of the SLIC are controllably optimized by monitoring the differential DC tip-ring loop current, via sense resistors in the output paths of respective tip and ring output amplifiers driving the tip and ring conductors of the wireline pair of interest. The sense resistors have values several orders in magnitude smaller than the values of the feedback resistors of the tip and ring output amplifiers.
The magnitude of a DC tip voltage applied to the tip conductor is determined by the product of the value of the tip amplifier""s feedback resistor and a controllably adjustable tip bias current injected therethrough. In a similar manner, the magnitude of a DC ring voltage applied to the ring conductor is determined by the product of the value of the ring amplifier""s feedback resistor and a controllably adjustable ring bias current injected through the ring amplifier""s feedback resistor. The values of these tip and ring DC bias currents are controllably established so as to provide a differential DC voltage Vtr between the tip and ring terminals having the required overhead voltage (relative to ground and to battery, respectively), and supply the necessary DC current required to bias the phone at the far end of the loop which, as noted above, may be more than several miles away.
In order to appropriately set the values of the tip and ring bias currents, the loop currents flowing through the tip and ring sense resistors are differentially coupled to a transimpedance circuit of the type employed in transimpedance stage of the SLIC transmission channel described in the above-referenced ""505 application may be employed. This transimpedance stage transforms the differentially sensed tip and ring input currents into a precise, single ended voltage that is applied to a transconductance amplifier stage.
The AC and DC components of the differentially sensed loop current are separated by an RC (resistor-capacitor) passband filter coupled with a transconductance amplifier stage to which a summation voltage from the transimpedance amplifier is applied. The DC voltage component across the passband filter capacitor is coupled to an absolute value circuit, which produces first and second current components representative of the absolute value of the dc component of the loop current, and scaled by a prescribed factor. The first current component is coupled to a first comparator circuit to which first and second (scaled) threshold currents ITH1 and ITH2 are applied. The second current component is coupled to a second comparator, to which a third threshold current ILIM is applied.
As will be described, the first current threshold ITH1 corresponds to a dc loop current greater than the leakage currents which may be encountered in the subscriber line, and associated with a transition in the operation of the phone from an on-hook, quiescent mode to an active off-hook, call (voice signal transmission) mode. The second current threshold ITH2 corresponds to a higher valued dc loop current associated with the completion of the transition in the operation of the phone from the on-hook mode to the active off-hook mode. The third threshold ILIM corresponds to an upper dc loop current threshold greater than the second loop current threshold ITH2, and associated with the upper end of the active off-hook mode.
In accordance with the operation of the first comparator, as long as the first current component produced by the absolute value circuit is less than the first threshold current ITH1, the comparator supplies a first output current I1=0 over a first current path. On is the other hand, if the first current component is equal to or greater than the first threshold, the value of the comparator""s first output current I1 is proportional to the difference between the first current component produced by the absolute value circuit and the first dc loop current threshold value.
A second current path from a first current mirror to the first comparator provides a second output current I2 as follows. If the first current I1 produced by the first comparator is less than the second threshold current value ITH2, then the second current I2 is equal to the sum of the first threshold current and the first current I1. However, if the first current I1 is equal to or greater than the second threshold current value, the second current I2 is equal to the sum of the first and second threshold currents ITH1 and ITH2.
A second port of the first current mirror is coupled to an external reference resistor (RDC). The mirrored voltage across the reference resistor is coupled to a unity gain buffer, the output of which controls the current applied to a second current mirror. This current is mirrored at a (1:1) current mirror port and a (2:1) current mirror port of the second current mirror. The (1:1) current mirror port is injected through the tip amplifier feedback resistor, so that a fixed tip bias voltage based on the second current I2 is applied to the tip conductor. An additional, a relatively small tip path overhead current from an auxiliary current source is supplied through the tip amplifier""s feedback resistor.
The (2:1) current mirror port of the second current mirror couples a second mirrored current to a common node point between a pair of equal valued resistors. One of these resistors is coupled to a virtual ground circuit; the other resistor is coupled to a power supply or battery terminal VBAT. A auxiliary current source also supplies a relatively small valued ring path DC bias current to this common node. In addition, this common node is coupled to receive a mirrored current from a third current mirror, a first current mirror port of which is coupled to receive a current 13 from the second comparator, to which the second current component from the absolute value circuit is coupled.
Similar to the node connection for the first threshold current value to the first comparator, if the second current component produced by the absolute value circuit is less than the third (upper limit) threshold current ILIM, the value of the third current I3 is set equal to zero. On the other hand, if the magnitude of the second current component from the absolute value circuit is equal to or greater than the upper limit threshold current value ILIM, the third current I3 produced by the second comparator is proportional to the difference between the second current component and the upper limit threshold current ILIM.
The virtual ground device has an associated current generator which mirrors a current through one of the pair of equal valued, common node-connected resistors through the feedback resistor of the ring output amplifier. As a result, the DC voltage drop across the ring amplifier""s feedback resistor varies in accordance with the mode of operation of the SLIC. When differentially combined with the DC voltage drop across the tip amplifier""s feedback resistor, the tip-ring output voltage Vtr varies in an optimum relationship to DC loop current.
In particular, as long as the DC loop current is less than the first dc loop current threshold ITH1, tip-ring voltage Vtr is equal to a first fixed offset voltage value plus ITH1*RDC below the battery voltage VBAT. Once the first dc current threshold ITH1 is reached, then as long as the loop current remains less than the second threshold, the voltage Vtr transitions along a non-zero slope segment from its value at the first threshold point. When the loop current reaches the second threshold point, Vtr acquires a second fixed value which is defined to provide a prescribed amount of signal amplitude headroom relative to the battery voltage in order to avoid clipping of the AC signal. When the loop current reaches the upper limit ILIM, the differential voltage Vtr rapidly drops to zero along a very steep slope.